This invention relates generally to switch-mode regulators in general and, more particularly, to low output voltage switching regulators with isolation.
Switch-mode regulators are widely used to supply power to electronic devices, such as in computers, printers, telecommunication equipment, and other devices. Such switch-mode regulators are available in variety of configurations for producing the desired output voltage or current from a source voltage with or without galvanic isolation. The former are also known as an isolated power converters and the later are called a non-isolated power converters.
One of the more challenging loads for power supplies are microprocessors. Because most of the microprocessors are implemented in complementary metal-oxide-semiconductor (CMOS) technology, the power dissipation of the microprocessor generally increases linearly with the clock frequency and to the square of the power supply voltage. There are three common techniques used to reduce power dissipation: power supply voltage reduction, selective clock speed reduction and reducing capacitive loading of internal nodes within the microprocessor. The first two techniques may be used in combination and could be controlled by circuit designer. Even a small reduction in power supply voltage makes a significant reduction in power dissipation. Also, if the clock is removed or significantly slowed in portions of the microprocessor not being used at any given time, very little power is dissipated in those portions and the overall power dissipated is significantly reduced.
However, these power savings techniques come at a cost. Power supply current can swing widelyxe2x80x94from hundreds of milli-amperes to over few tens amperes with the microprocessor unable to tolerate more than a few percent change in voltage. Further, the change in current can occur in tens of nanoseconds and may change in an order of magnitude. The power supply designed to supply the microprocessor must have a sufficient low impedance and tight regulation to supply such dynamic power consumption. With output voltages approaching 1V or even sub-volt levels and load currents approaching hundred amperes, the power supplies are very difficult to make and control and still operate efficiently.
In addition, a dedicated power supply for the microprocessor has to be placed in close proximity to the microprocessor. Thus, the power supply must be small and efficient. To meet these requirements, a small DC-to-DC switching power regulator is usually used. The widely used switching regulator to convert a higher input voltage (usually 5V or 12V) to a lower output voltage level is xe2x80x9cbuckxe2x80x9d regulator. In applications where input voltage, often referred as bus voltage, is greater than 12V (e.g. 24V or 48V) single non-isolated switching regulator, such as xe2x80x9cbuckxe2x80x9d regulator, becomes very difficult to make small and efficient. In addition, in these applications a galvanic isolation is very often required thus switching regulator needs to have isolation. One of the most common approaches is to use two stage conversion. First stage conversion is provided using an isolated switching regulator in order to provide galvanic isolation and to step-down high voltage input bus (typically 48V) to lower voltage bus (5V or 3.3V). The second stage is then realized using xe2x80x9cbuckxe2x80x9d switching regulator. Obvious disadvantages of this approach are need for two switching regulators which increases overall cost and reduces overall efficiency.
Three kinds of feedback are generally used to control the operation of the regulator: voltage alone (with current limiting), voltage with peak current control, and voltage with the average current control. For reference see xe2x80x9cFueling the Megaprocessorsxe2x80x94Empowering Dynamic Energy Managementxe2x80x9d by Bob Mammano, published by the Unitrode Corporation, 1996. The voltage with the average current control type of regulation is generally preferred over the other types for the described reasons. Regardless of which type of the feedback control is used, there is need for output current sensing either directly on indirectly. The most common approach is to the sensing resistor in series with the output inductor.
The circuit reconstructs the output inductor current as a differential voltage across the sensing resistor. Most integrated circuits using this approach regulates output voltage with current mode control and use the signal for output voltage feedback. The sensing resistor value must be on one side large enough to provide a sufficiently high voltage, usually tens of millivolts, to overcome input offset errors of the sense amplifier coupled to the sensing resistor and yet small enough to avoid excessive power dissipation. Since the power dissipated in the sensing resistor increases with the square of current, this approach has the obvious efficiency drawback with high output current and low output voltage. For low voltage, high current applications, the value of the sensing resistor may be close or even higher than the on resistance of the power switch and inductor which are minimized for maximum efficiency. Thus, sensed signal is relatively small and requires use of more expensive either comparators or amplifiers. Further, the circuitry implementing the average current control technique is significantly more complicated than the circuitry of the other two techniques.
Power inductors are known to have parasitic (or inherent) winding resistance, and therefore can be represented by an equivalent circuit of a series combination of an ideal inductor and a resistor. When direct (DC) current flows through the inductor (or a current having a DC component), a DC voltage drop is imposed across the inductor, which voltage is a product of the magnitude of the DC (component of the) current and the parasitic resistance of the inductor. Since such an inductor may already be present in the circuit, there is no an additional loss of efficiency in using the inductor for this purpose.
Parasitic resistance of the output inductor is used for current sensing as described in U.S. Pat. No. 5,465,201, issued to Cohen, U.S. Pat. No. 5,877,611, issued to Brkovic, U.S. Pat. No. 5,982,160, issued to Walters et al. and U.S. Pat. No. 6,127,814, issued to Goder, all of which patents are hereby incorporated herein by reference. In U.S. Pat. No. 5,877,611 and U.S. Pat. No. 5,982,160 load current dependant output voltage regulation employing inductor current sensing is proposed. Again, sensed signal is limited to product of inductor""s winding resistance and inductor current and can be increased only by means of active amplification, which adds complexity, inaccuracy and mostly additional cost. In order to maximize efficiency of the converter, inductor""s parasitic resistance (particular at high current applications) has to be minimized thus, the sensed signal is relatively small and requires use of more expensive either comparators or amplifiers. Very often error due to offset in comparator and/or amplifier is larger than variation in the winding resistance of the inductor (windings printed on the PCB).
Perhaps the most common approach to sensing the output inductor current indirectly in isolated topologies is to use sense resistor in series with power switches or current sense transformer. Use of sense resistor in single ended topologies, such as for example forward, flayback and others, as well as in full-bridge and push-pull topologies, allows that one end of sense resistor is coupled to GND pin of control chip, usually coupled to input return, which simplifies current sensing. On other hand, the sensing resistor value must be large enough to keep the sensed signal above the noise floor and yet small enough to avoid excessive power dissipation. In case of half-bridge converter, for example, this approach is not good since only one power switch is coupled to input return and sensed signal does not reflect current through second, floating power switch. Using sense resistor in return input path is also not good solution since sensed current is not exactly current through primary side switches but rather an input current of the converter smoothed by input capacitors. Also, sensed switch current differs from the output inductor current due to magnetizing current of isolation transformer which also varies with the input voltage. Using current sense transformer is not practical solution for two main reasons: it still measures the sum of the magnetizing and reflected output inductor current and it becomes difficult to implement in low profile high power density switching regulators.
Therefore, it is an object of the present invention to provide an efficient isolated switching regulator having a voltage and current control technique.
It is another aspect of the invention to provide a switching regulator having a fast transient response with relatively simple control circuitry.
It is a further aspect of the invention to provide a switching regulator design that allows for parallel operation.
This and other aspects of the invention may be obtained generally in a computing system, a switching regulator for powering a load including microprocessor, the switching regulator comprising an input voltage source, a switching circuit and an transformer for coupling the input voltage source and an output stage, the switching circuit comprising at least two controllable power switches and a first and a second rectifier switch, the transformer comprising at least one primary winding coupled to the input voltage source via at least two controllable power switches and a first and a second secondary windings, the first and the second secondary winding having inherent resistances, the first and the second secondary windings coupled together in series at a first node, the first and the second rectifier switches coupled in series at a second node, the first secondary windings coupled to the first rectifier switch at a third node, the second secondary winding coupled to the second rectifier switch at fourth node, the output stage having an output stage input and a output stage output, the output stage input coupled to the first node and the second node, the output stage output coupled to a load circuit, the output stage input and the output stage output having a common node, the output stage comprising an inductor for providing output current to the load circuit, the inductor having a first terminal for coupling to the output stage input and a second terminal for coupling to the output stage output, a first resistor having a first and a second terminal, a second resistor having a first and a second terminal, a capacitor having a first and a second terminal, the first terminal of the first resistor coupled to the node three and the first terminal of the second resistor coupled to the node four, the second terminal of the first resistor coupled to the second terminal of the second resistor and the first terminal of the capacitor at a node five, the second terminal of the capacitor coupled to the second terminal of the inductor and the node five is coupled to an input of an error amplifier for controlling the switching circuit.
In one embodiment of the present invention a third resistor is disposed in parallel with the capacitor for additional adjustment of the droop voltage.
In one embodiment of the present invention a first impedance is coupled between the fifth node and an inverting input of the error amplifier and a second impedance is disposed between the output and inverting input of the error amplifier.
In one embodiment of the present invention a fourth resistor is connected to the error amplifier inverting input for adjusting output voltage to a reference voltage.
In one embodiment of the present invention a third impedance is coupled between the output stage output and inverting input of said error amplifier. In one embodiment, the first impedance provides DC coupling so the output voltage is load dependant while, the third impedance provides only AC coupling from the output voltage of the switching regulator for improved transient response. In yet another embodiment, the first impedance provides only AC coupling while, the third impedance provides DC coupling from the output voltage of the switching regulator for load independent regulation. By selecting the first impedance to provide only AC coupling, current feedback is fed into the error amplifier only during the load current transients thus providing more stable transient response.
In one embodiment of the present invention a switching regulator with load dependent regulation is parallel-coupled to a plurality of switching regulators sharing a common reference voltage and common output coupled to the load.